Principle of electrically small antennas

According to the definition, the size of an ESA is much smaller compared to the wavelength $\lambda$ and can be defined as [Reference Wheeler22]:

(1)\begin{equation}ka = \frac{2\pi}{\lambda} a \lt 1,\end{equation}

where $a$ is the radius of a sphere (also called the radiansphere) enclosing the antenna, and $k$ is the wavenumber. Based on (1), the equation defining the fractional bandwidth ( $B_\mathrm{frac}$) of the antenna is:

(2)\begin{equation}B_\mathrm{frac}=\frac{S-1}{Q_\mathrm{min}\sqrt{S}},\end{equation}

where $S$ is the standing wave ratio and the minimum $Q$ value ( $Q_\mathrm{min}$) is:

(3)\begin{equation}Q_\mathrm{min} = (ka)^{-3}+(ka)^{-1}.\end{equation}

This expression is the well-known Chu–Harrington limit [Reference Chu21, Reference Hansen23, Reference McLean24].

In this work, a challenging set of limits has been defined for the antenna due to the compactness of the device, as shown in Fig. 1. Different spherical radii calculated using (1)-(3) are visualized in the two-dimensional $xy$ plane, and the area allocated for the antenna placement is highlighted in pink. Note that the vertical area of the spheres cannot be utilized due to the geometrical constraints of the device. On the other hand, the expected fractional bandwidth ( $Bw$) limitations for the radii of Fig. 1 are plotted in Fig. 2. When estimating the $Bw$, the low band frequency is calculated based on $S_{lb}$ = 3 at $f$ = 830 MHz, and the high band case uses $S_{hb}$ = 2 at $f$ = 1.95 GHz. The plotted red and black lines in this figure highlight the electrically small regions for the high and low bands, respectively. It is worth noting that the ESA limit in the high band is at $a$ = 24 mm, which is slightly smaller than the size of the design. Despite not being considered electrically small at the high band (with $ka_{hb}$ = 1.35), a contrary and challenging condition exists at the low band (with $ka_{lb}$ = 0.57).

Figure 1.

Visualization of the different enclosing spheres with radius $a$ according to (1). Area allocated for the antenna design is highlighted in pink. The major dimensions of the PCB are annotated.

Figure 2.

ESA bandwidth limits calculated using eq. (1) in comparison to the enclosing sphere radius $a$. The markers indicate the following cases: I) Size required for high-band operation: $a$ = 21 mm / $Bw$ = 26%. II) Size of the design: $a$ = 33 mm / $Bw$ = 16%. III) Size required for low band operation: $a$ = 43 mm / $Bw$ = 31%.

Due to the narrow bandwidth available from ESAs, achieving instantaneous and wide bandwidth is very challenging. To widen bandwidth, multiple resonating structures that operate in complementary, closely-spaced bands could be implemented, similar to the technique in [Reference Koutinos, Zekios, Georgakopoulos and Kyriacou25]. However, due to the limited space available in this device, such a technique is not a viable option. Instead, enabling multiband operation using a switching element is a more realistic alternative. Switches do not require a large footprint, but enable the switched operation of the antenna resulting from multiple matching networks [Reference Garcia, Andújar, Pijoan and Anguera17]. This, in turn, allows the antenna to operate over the wider targeted bandwidth of operation when switching across closely-spaced bands. In this paper, the number of states is kept small (two) to keep the control circuitry manageable.

In practice, the band-switchable antenna system is a concept intended for the Bittium Faros^TM series of wearable cardiac monitors [26]. This requires the antenna section to be accommodated in an area of 14 $\times$25 mm $^2$ (0.04 $\times$0.07 $\lambda^2$) (area highlighted in pink). Besides that, it is assumed that the feed line and lumped components will be integrated within the same area as other circuits (printed circuit board areas in white in Fig. 1). These aspects will be explained in more detail in the following subsections.

Characteristic mode analysis

Characteristic mode analysis is an efficient tool to analyze the potential performance of an antenna design. It provides insight into the possible modes excited on a structure at a given frequency, allowing a more analytical path to identify the most suitable feed locations [Reference Martens, Safin and Manteuffel27, Reference Cabedo-Fabres, Antonino-Daviu, Valero-Nogueira and Bataller28].

To have an initial idea of the potentially available characteristic modes from the structure based on the lowest targeted frequency of 900 MHz, a series of characteristic mode analyses is first performed with varying ground plane sizes. When the ground plane is scaled to 150, 200, and 250% of the nominal size, the size of the antenna element remains constant in all cases. The modal significance of the first four characteristic modes at 900 MHz, when the ground plane size is increased in both axial dimensions (x and y), is shown in Fig. 3. It can be observed that the set physical size limitation (100%) of the device significantly restricts the availability of significant modes for antenna radiation. As the device size is increased, more modes become significant at the lower frequencies, with selected modes retaining their operation at the 1.7 GHz band.

Figure 3.

The change in modal significance of the first four characteristic modes at 900 MHz: a) Mode 1, b) Mode 2, c) Mode 3, and d) Mode 4, when the ground plane size is increased. The size of the ground plane (white) in Figure 1, was increased by 50% (red), 100% (green), and 150% (magenta), in both the x and y directions, whereas (blue) is the reference size. The different mode currents can be seen in Figure 4.

Next, based on the 100% ground plane size limitation (blue line in Fig. 3), the corresponding surface currents generated by different modes are investigated at 900 MHz. The results are illustrated in Fig. 4. Mode 1 in Fig. 4(a) has current flowing at the edges of the major axis, whereas for Mode 2 shown in Fig. 4(b), the main current is distributed evenly and directed along the minor axis. On the other hand, the currents of Mode 3 depicted in Fig. 4(c) are also directed along the minor axis, but the flow is mainly at the edges of the ground plane. The fourth highest modal significance, mode 4 in Fig. 4(d), has no current in the ground plane and a relatively high activation frequency. Based on this, it can be concluded that the antenna element as a standalone structure does not produce significant radiation performance for this design, but requires the ground plane to be excited to achieve reasonable performance. This kind of operation is similar to capacitive coupling element (CCE) based handset antennas [Reference Vainikainen, Ollikainen, Kivekäs and Kelander29]. Due to the strict size requirements of the device, the design is kept in the set limits and not increased from 63.2 $\times$25 mm $^2$.

Figure 4.

The surface currents at 900 MHz for the four modes shown in Figure 3: a) Mode 1, b) Mode 2, c) Mode 3, and d) Mode 4. The orientation is similar to the model in Figure 1.

Figure 5 shows the current distribution for three different frequencies. From these, we can analyze how the different parts of the design affect the performance of the design. In Fig. 5(a), it can be seen that there is a high current in the shorted section. As seen in Fig. 4(a), the strongest characteristic mode also has the same strong current in the shorted strip. There is also a relatively high current in the meander line that reduces as the frequency increases.

Figure 5.

Maximum current distribution at a) 936 MHz, b) 1.76 GHz, and c) 2.06 GHz. Both sides of the antenna are shown.

The current in the ground plane stays in the same range regardless of the frequency, with slight deviations as the characteristic modes change. For the 1.76 GHz current distribution, the horizontal section before the meander line and the meander line itself have higher current flows compared to the 2.56 GHz case. These are also important for the operation at the high band. The actual antenna element has quite a stable excitation through all the different frequencies. As a conclusion, all of the parts from the design play a crucial role in the achieved performance, with the ground plane being the limiting factor for achieving a lower frequency.

Antenna design and aperture tuning

The antenna design is modified from the design presented in [Reference Ban, Liu, Chen, Li and Kang19] with more challenging space requirements. The unique goal is to retain most of the bandwidth in the LB and maintain the operation in the HB despite this space limitation. The design consists of a U-shaped feeding strip and two sections of shorting strips. The two sections of the shorting strips are connected using an inductor. The first part of this shorting strip is placed on the same PCB layer as the feeding strip, whereas its second part is bent in a meandered form inside the 3D-printed casing of the device. To enable more precise impedance tuning, the antenna aperture is tuned by an inductor (denoted by circle no 2), as shown in Fig. 6. The grounded meander line acts as an extension to the ground plane of the PCB, having the electrical length extended with the aperture tuning inductor [Reference Ali and Payandehjoo30]. Such method maintains design simplicity and avoids the need for inductance tuning using complex geometrical modifications due to space limitations.

Figure 6.

Meanderline structure with dimensions. 1) The feed location and 2) the aperture tuning component location.

Antenna simulations and optimization are performed using CST Microwave Studio Suite 2023, whereas optimization for the matching circuit is performed using Optenni Lab software [31]. The reconfigurable antenna system is designed on a cost-efficient, 1-mm thick FR4 substrate ( $\varepsilon_\mathrm{r}$ = 4.3 and metal thickness of 18 $\unicode{x03BC}$m). The PCB is housed in a 3D-printed case, with an area of 66 $\times$33 mm $^2$ and which lifts the PCB 10 mm from the bottom to allow placing the meander line below the PCB.

Antenna operation

The initial design consists of a patch-like structure, feed position, and the grounded segment is denoted with 1 and 2, respectively, in Fig. 7. This provides the basis for the high-band operation of the structure, with the grounded segment being observed to significantly contribute to this behavior. The effect of the ground extension and aperture tuning from Fig. 7 is shown in Fig. 8.

Figure 7.

The different radiating structures, for which $|S_\mathrm{11}|$ results are shown in Figure 8: a) Grounded strip and simple patch, b) added meander line, c) modified patch, and d) additional aperture tuning inductor (2.2 nH, same as in the final structure of Figure 6). Here, (1) marks the feed port, (2) marks the grounding, and (3) marks the placement of the aperture tuning inductor.

Figure 8.

The effect of the different antenna parts on the input reflection coefficient $\left(|S_{11}|\right)$. The different design stages are visualized in Figure 7. The parts are not optimized for free-space operation; thus, the actual performance in this figure should only be regarded as a comparison between the accompanying results.

When adding the meander line structure, which extends below the ground plane of the PCB, as shown in Fig. 7(b), a new resonance appears at the low band. This modification also slightly boosts the high-band operation. In Fig. 7(c), the patch is modified to its final form by introducing a T-shaped cutout and moving the grounding point of the shorted strip. With this modification, the high-band performance is again slightly improved, and the low-band resonance shifts down in frequency.

To further tune the low-band operation, a discrete inductance (2.2 nH) is added between the grounded segment and the meander line. This shifts the low-band resonance toward the target range, with the amount of frequency shift defined by the additional inductance value. It can be observed that the higher the inductance value, the lower the frequency shifts. However, this comes at the cost of decreased total efficiency, and a proper evaluation of the trade-offs is required.

Antenna input matching

To further improve the antenna input impedance matching (to 50 $\Omega$) in the low band, lumped matching elements are used due to their compact size compared with that of the antenna. Besides that, these components also allow frequency tunability with a switching network.

The radiating structure is shown in Fig. 9. It is designed in a switchable format to operate in two segments in the low band, using Configurations 1 (875–900 MHz) and 2 (898–934 MHz). In Fig. 10, the RF input is fed by an SMPM connector (Rosenberger) to a switched matching network. This network is switched between the two configurations using a simple combination of an inverter (Rohm) and a mechanical switch (Wurth) as a proof-of-concept. The matching networks are located between RF switches (Qorvo). The matching network for Configuration 1 consists of a parallel capacitor (Murata 5.7 pF) and a series inductor (Murata 6 nH). For Configuration 2, a simple series capacitor (Murata 3.3 pF) is used.

Figure 9.

Top view of the antenna element with dimensions annotated in mm. (1) Marks the feed location, (2) marks the grounding, and (3) the aperture tuning component location.

Figure 10.

Schematic of the matching- and control-circuit, with the used components annotated.

Measurement setup and body phantom

The optimized design was then fabricated, as shown in Fig. 11. $S$-parameter measurements were performed using an HP 8720ES vector network analyzer (VNA), whereas radiation patterns and efficiency were assessed using a SATIMO Starlab measurement system, as seen in Fig. 12. $S$-parameter measurements were performed on a homogeneous body phantom whose dimensions are 18 $\times$24 $\times$15 cm $^3$. The size of the phantom was then reduced to 10 $\times$20 $\times$8 cm $^3$ for the measurements of radiation pattern and efficiency due to the space and weight constraints on the antenna measurement platform. The simulated phantom shares the dielectric properties of the Basic phantom design of [32], and the phantom for measurements was made to match the same model using the recipe/part of the same recipe as the body model in [Reference Särestöniemi, Singh, Dessai, Heredia, Myllymäki and Myllylä33]. An RF cable and an SMPM-to-SMA adapter are used throughout the measurements.

Figure 11.

Close-up of the fabricated design, with the matching circuit and RF input.

Figure 12.

The reduced body phantom (10 $\times$20 $\times$8 cm $^3$) setup in the Satimo Starlab, for efficiency and gain measurements.

On-phantom measurements were performed using a 10 mm thick piece of Rohacell foam spacer, as seen in Fig. 13. The spacer is emulating the on-body adapter, which is included in the Bittium Faros^TM device [26]. Under the spacer, a piece of copper tape was used in on-body results as a reflector, as seen in Fig. 14, to reduce SAR, with a size roughly matching the size of the device [Reference Paracha, Rahim, Soh and Khalily20].

Figure 13.

On-body $S$-parameter measurements, with a 18 $\times$24 $\times$15 cm $^3$ body phantom.

Figure 14.

The Rohacell spacer and copper tape reflector next to the antenna.

Free space performance

Scattering parameters

Figure 15 presents the simulated and measured free space reflection coefficient for the antenna system. There is a slight shift upward in the measured resonant frequency in the low band and significant dampening in the matching compared to simulations. The high band displays an opposite behavior, with the measurement having a higher peak in the matching and the resonance in a slightly lower frequency, although the high band retains the required bandwidth.\RightSideStrip

Figure 15.

Simulated and measured reflection coefficient of the antenna in Free space. Here, Conf1 and Conf2 refer to the different matching network configurations.

Efficiency

Figure 16 shows the total efficiency of the antenna in free space. The efficiency peaks are around 0.9 GHz ( $-$4 dB), 0.93 GHz ( $-$3 dB), and 1.9 GHz ( $-$2 dB), which align well with the deepest resonances in the reflection coefficient of Fig. 15. A higher efficiency would be preferable, but ESAs have naturally low efficiency [Reference Nepa and Rogier7]. Although the measured reflection coefficient has a deeper notch at the high band compared to the simulation, the efficiency does not exhibit the same behavior. This is likely due to additional losses in actual components, which lower the peak efficiency of the prototype.

Figure 16.

Simulated and measured free space total efficiency of the antenna.

Radiation patterns

In Fig. 17, the simulated and measured gain is plotted for the free space case. The low band gain is slightly below 0 dB with deviation to as low as -10 dB, with a mostly omnidirectional pattern. The high band is similar with a deviation at $\phi=90^{\circ}$ and $\theta=120^{\circ}$ in Fig. 17(d).

Figure 17.

Simulated (dashed) and measured (solid) free-space radiation patterns at: (a) 920 MHz, $\phi$ = 0 $^{\circ}$ cut, (c) 920 MHz, $\phi$ = 90 $^{\circ}$ cut, (b) 1.9 GHz, $\phi$ = 0 $^{\circ}$ cut, and (d) 1.9 GHz, $\phi$ = 90 $^{\circ}$ cut. The orientation of the antenna can be seen in Figures 1 and 11. a) 920 MHz / $\phi=0^\circ$. b) 1.9 GHz / $\phi=0^\circ$. c) 920 MHz / $\phi=90^\circ$. d) 1.9 GHz / $\phi=90^\circ$.

On-body performance

This subsection describes the on-body measurements and their results. The body phantom is described in Section III-A. with the general measurement setup.

Scattering parameters

The simulated and measured on-body reflection coefficients are shown in Fig. 18 for both switch configurations. The combined $-$6 dB bandwidth are from 875 MHz to 934 MHz in the low band, and from 1.7 GHz to 2.2 GHz in the high band The simulated low-band resonances are clearer (with significantly deeper dips) and a gap between the two configurations. In the high band, there is a more noticeable difference between the simulated and measured results relative to their performance in free space. Although both curves exhibit a dual-resonance response when operated in proximity to the body phantom, a larger gap between the two resonance peaks is observed in simulations.

Figure 18.

Comparison of simulated and measured on-body reflection coefficients. Bandwidth is defined according to the –6-dB matching criterion (indicated by the black dotted line).

The measured on-body bandwidth in the low band (875–934 MHz) is slightly wider than the simulated one (886–910 MHz) and covers the entire target frequency range. In the high band, the measured on-body results featured a bandwidth of 1714–2190 MHz, which almost satisfies the required bandwidth specifications. A larger discrepancy is seen in the high band due to the practical differences in the properties (relative permittivity and conductivity) of the phantoms across frequency. The properties of the phantom model used in simulations are extrapolated across the frequency of interest, whereas the properties of the fabricated phantom can be made close to the required values using the current manufacturing method in either the low or high band only. In this case, considering the more sensitive behavior of the antenna performance in the low band due to the larger distance needed to decouple them, the phantom has been fabricated based on the properties in this (low) band to evaluate the worst-case performance scenario of on-body coupling. Besides that, unlike in simulations, it is challenging to ensure that the fabricated phantom produces exact same homogeneous permittivity and conductivity in practice. Variation of these properties across the phantom’s width and length may occur and affect the accuracy of the measurement results. Despite these uncertainties, it is observed that the simulated and measured $|S_\mathrm{11}|$ in the high band exhibited similar bandwidths, with differences mainly observed in the depths of the resonances.

In Fig. 19(a), the low band curves show that the simulated results are overcoupled, while the measured results are undercoupled. On the other hand, in Fig. 19(b), the simulated high band results are undercoupled, whereas the measured results are almost critically coupled. When considering the high band results, the two simulated resonances of Fig. 18 are clearly visible as separate loops, whereas the measurements show a nascent dual resonance. Based on these observations, moving the impedance at the two resonances closer to the center of the Smith chart allows improvement of the overall matching level across the high band.

Figure 19.

Comparison of simulated and measured complex reflection coefficients. a) Low band: 0.6–1.2 GHz. b) High band: 1.5–2.5 GHz.

Efficiency

The efficiency simulation uses the smaller body phantom due to the reasons mentioned in Section III-A. The total efficiency of an antenna refers to the power fed to it compared to the power radiated, accounting for all losses along the signal path. In this work, a total efficiency of at least $-$10 dB is desired throughout the operating frequency band. Results illustrated in Fig. 20 in both the low and high bands indicate that the total efficiencies correspond well to the reflection coefficient results, and the $-$10 dB target efficiency is achieved over the entire $-$6 dB bandwidth.

Figure 20.

Comparison of simulated and measured on-body total efficiency. The black line marks the –10 dB line used to determine the efficiency bandwidth.

Compared to the free space efficiency in Fig. 16, only a drop of $-$2 dB is observed when the body phantom is introduced. This effect is contributed by the existence of the reflecting copper plate, and is also comparable to the study in [Reference Hall, Hao, Nechayev, Alomainy, Constantinou, Parini, Kamarudin, Salim, Hee, Dubrovka, Owadally, Song, Serra, Nepa, Gallo and Bozzetti8]. The difference between the low band efficiencies is a result of the lower frequency that affects the electrical size and radiation characteristics of the antenna.

Radiation patterns

The current ESA system is intended to be used on the body to collect ECG-based health information and channel it to a base station. Therefore, in this application, the antenna radiation patterns need to be designed appropriately, with the main beam pointing away from the user. At the same time, the existence of the ground plane will ensure that the performance of the antenna is not easily influenced by the coupling to the body. Moreover, high back radiation can increase SAR levels, and this should be avoided.

Prior to the on-body radiation pattern measurement, the size of the body phantom was reduced (from 18 $\times$24 $\times$15 cm $^3$ to 10 $\times$20 $\times$8 cm $^3$) to comply with the mechanical rotation of the StarLab measurement system. Nonetheless, it is expected that this change will have minimal effects in evaluating the true performance of the antenna on the body, as the distance from the edges of the antenna to the edges of the body model is at least 0.1 $\lambda$. In Fig. 21, the gain patterns for the design are shown, for both the low band (920 MHz) and the high band (1.9 GHz), on two different axes. The gain toward the body is significantly increased in measurements in the low band, as seen in Fig. 21(a) and (b). In other aspects, the measured gain is similar to the simulated gain. Gain simulations were also performed using the smaller body phantom model.

Figure 21.

Simulated (dashed) and measured (solid) on-body radiation patterns at: (a) 920 MHz, $\phi$ = 0 $^{\circ}$ cut, (c) 920 MHz, $\phi$ = 90 $^{\circ}$ cut, (b) 1.9 GHz, $\phi$ = 0 $^{\circ}$ cut, and (d) 1.9 GHz, $\phi$ = 90 $^{\circ}$ cut. The orientation of the antenna can be seen in Figures 1 and 11. a) 920 MHz / $\phi=0^\circ$. b) 1.9 GHz / $\phi=0^\circ$. c) 920 MHz / $\phi=90^\circ$. d) 1.9 GHz / $\phi=90^\circ$.

Specific absorption rate

The SAR is a measure of power absorption in body tissue/phantom, and its limit is 1.6 W/kg averaged over a 1-g mass of tissue according to the Federal Communications Commission (FCC) standards in the United States [3]. This regulation is chosen as it is one of the strictest. In comparison, the IEEE limit is 2 W/kg averaged over 10 g of tissue, which is relatively easier to obtain.

In this work, the SAR for the band-switchable ESA system is simulated in CST using a homogeneous body phantom. An accepted power of 23 dBm was used to roughly match a possible use case power. A 10-mm gap is chosen to emulate the spacing between the antenna and human body in an actual use case, which includes the ECG electrode and an adapter for the device. The effect of the additional metal layer is evaluated in terms of reflection coefficients in Fig. 22. In the low band, a stronger resonance peak is seen for Configuration 1, without any obvious changes in Configuration 2. On the other hand, in the high band, a similar increase in resonance strength is noticed for the resonance peaks, resulting in a slight degradation of the reflection coefficient in the center of the whole high band. Despite this, the measured $|S_\mathrm{11}|$ of the prototype with the phantom in the high band indicated satisfactory performance while producing SAR values within the accepted regulatory level.

Figure 22.

Simulated reflection coefficient with (Sim) and without (NC) the SAR reduction layer. Both of the results employ a 10 mm gap between the model and the phantom.

The SAR results are shown in Table 1, indicating satisfactory compliance with the safety levels. These SAR levels can be further reduced by modifications of the transmit power duty cycle [Reference Zradziński, Karpowicz and Gryz35]. Besides that, future research could include more advanced and compact metamaterial-based solutions such as artificial magnetic conductor (AMC) reflectors [Reference Atrash, Abdalla and Elhennawy36–Reference il Kwak, Sim, Kwon and Yoon38] to reduce SAR levels by limiting backward radiation. However, such solutions come at the cost of a larger area requirement for their implementation, as they are typically designed in the form of an array of unit cells at their operating frequency. Considering the need for such a reflector in the low band in this case, a significantly compact AMC reflector is needed to ensure full compliance with the current operation of the health monitoring device in practice. This is certainly a challenging aspect that can be pursued in the next stage of the research. It should be noted that SAR measurements were not possible at the time of the investigation. However, considering the reasonably large safety margins from the SAR limits, it can be expected that the measured values would also fall well within the acceptable limits [Reference Soh, Vandenbosch, Wee, van den Bosch, Martinez-Vazquez and Schreurs39].

Table 1.

Simulated SAR results with an accepted power of 23 dBm and when averaging over a 1-g mass of tissue

Comparison with the state-of-the-art

In the literature, there are only a few similar works on ESA design for on-body operation, which also include measurements. Nonetheless, several of the most relevant works are compared in Table 2. In [Reference Ruaro, Thaysen and Jakobsen13], an ESA working at 2.4 GHz is introduced for hearing aid applications. The research does not focus on bandwidth enhancement, but rather on improving radiation efficiency. In [Reference Orefice, Pirinoli and Dassano14], an ESA designed to operate in the sub-1 GHz band is proposed. The size is significantly larger, and the investigation is focused on the wearable aspect and radiation patterns. On the other hand, in [Reference Virushabadoss, Quaye and Henderson15], two designs for operation below 400 MHz are proposed. Both designs are remarkably compact compared to their frequency. However, this significantly affects the achieved bandwidth. Neither of the designs is meant for on-body use, but as a study into ESAs. Next, in [Reference Tang, Lin and Ziolkowski11], a Yagi-based antenna with polarization reconfigurability is introduced. This design utilizes the vertical space in the enclosing ESA sphere with a stacked design. In [Reference Sambandam, Kanagasabai, Mohammed, Thirunavukkarasu, Shanmuganathan, Ramadoss and Palaniswamy34], a small wearable antenna is introduced. The achieved footprint is small, and the targeted frequency bands are narrow, but sufficient. This design also achieves good efficiency. In [Reference Garcia, Andújar, Pijoan and Anguera17], a reconfigurable antenna system is proposed, operating from 698–960 MHz and from 1710–2170 MHz. This is achieved using a seven-state switchable matching network, enabling wide bandwidths in both frequency bands of interest with decent efficiency. However, as this method relies heavily on the ground plane, it may not be practical to achieve the targets in this work, besides not being intended for on-body use. Perhaps the most relevant comparison for this work is from [Reference Hajj, Person and Wiart16], in which a rectangular-spiral dual-band antenna operating in similar frequencies is proposed. Moreover, the effect of the proximity to the body is studied. A major difference between that work and the proposed antenna lies in the antenna-to-ground plane size ratio, which is more restricted in this work.

Table 2.

Comparison with other relevant work

¹ With a 150×140mm² ground plane.

² –6dB bandwidth, compared with –10dB bandwidth.

³ The high band cannot be considered as electrically small.

⁴ With 12mm to the phantom.

⁵ Averaged total efficiency.

⁶ Total efficiency.

⁷ Two bands at higher frequencies not reported here.