System design

Figure 1 shows the main functional parts of the receiving FE module as a block diagram.

Fig. 1.

Block diagram of the mm-wave FE with two steerable beams (beam #1 and beam #2 fed by the upper and lower parts of the circuit, respectively) and controllable polarization (H- and V-polarizations created in the left-hand and right-hand parts of the circuit, respectively). The beamformers and down-converters of beam #1 are shown explicitly, whereas the control circuits for beam #2 are indicated in a simplified way.

Low-noise amplifiers (LNA) decouple the dual-polarized antenna signals from the beam splitting power dividers. Phase shifters perform the beam steering, separately for each of the two orthogonal linear polarizations. The combined signals are down-converted by image reject mixers with internal local-oscillator (LO) frequency doubling, amplified, and summed with the orthogonally polarized component by a vector-multiplier integrated circuit. The weighted summation allows for an arbitrary rotation of the resulting polarization. The weights are set by DC voltages, because the multiplier is capable of handling baseband input signals. These functionalities are doubled and fed to a second beam, thus generating two separate agile beams. A common LO signal is frequency-doubled and power divided and supplied to all four mixers. All transmission paths are phase matched and allow an uncalibrated operation. The intermediate-frequency (IF) circuits and the integrated four-element patch antenna array currently limit the overall bandwidth to ~400 MHz, whereas the mm-wave block spans the entire frequency range from 28 to 31 GHz. Additional circuit boards deliver analog and digital control signals. The FE was assembled in the multi-layer low-temperature co-fired ceramics (LTCC) technology. Figure 2 displays a photograph of the assembled module, which consists of nine layers DuPont™ 9K7.

Fig. 2.

Photograph of the assembled FE with SMP plugs for RF connections, and a power supply and control connector. The module size measures 74 mm × 74 mm [Reference Spira, Blau, Thomä and Hein14].

Although previous study dealt with the comprehensive evaluation of the analog mm-wave transmission properties, this paper describes the results of recent OTA testing in a system context in greater detail, additionally addressing antenna design and measurements, because the antenna parameters determine to a large extent the performance of the overall system and thus the link-level parameters of broadband digitally modulated waveforms received with this FE, similar to realistic signals relevant for 5G mobile communications.

Antenna design

Originally, a 4 × 4 antenna array was envisaged. The active components required for such an array size, especially pre-amplifiers and active mixers, which necessarily have to be located near the antenna feed points, induce a dissipation power density exceeding the thermal limit; moreover, the available circuit area is insufficient in this case.

Since polarization agility should be ensured from the outset, in addition to beam steering, dual-polarized antenna elements have to be used. With 4 × 4 antenna elements, this means that the required LNAs, phase shifters, and mixers are doubled, so that high-power dissipation can be generated in a small area or volume. The resulting heat cannot be easily dissipated using conventional means like convection cooling using a heat sink. In addition, the size of the Green Tape Material DuPont™ 9K7 was limited to 75 mm × 75 mm of the processing area, which was a further restriction for the antenna design.

The basic idea of separately processing the linear H- and V-components of the polarization of the incident antenna signal is provided in Fig. 1. This also ensures polarization agility for elliptical polarization. In order to avoid the high thermal power density and taking the given area limitation into account, it was ultimately only possible to implement one antenna line consisting of four dual-polarized antenna elements in the design presented here.

Due to the decision to realize the demonstrator in LTCC technology and to operate it in the 28–31 GHz range, it was decided to adapt the antenna design to these conditions. In order to achieve a sufficiently broadband and dual-polarized antenna element with as little efforts as possible, and accounting for the technological LTCC limitations in terms of layer thickness and dimension tolerances, the choice fell on a square-shaped microstrip patch element, which basically can excite two orthogonal linear polarizations.

Due to the high permittivity of 9K7 (ɛ_r = 7.07), with a layer thicknesses of 0.315 mm (according to LTCC design rules), a correspondingly lower bandwidth of a microstrip patch must be expected if the excited field volume is completely filled with the ceramic material. For an estimation of the expected percentage bandwidth, equation (1) according to [Reference Kumar and Ray15] can be used:

(1)$$\% BW = A \times \displaystyle{{\;h} \over {{\it \lambda} _0\;\sqrt {\varepsilon _r} }} \times \;\sqrt {\displaystyle{W \over L}} \;$$

where the parameter A takes different values depending on the effective electrical thickness h_eff = h/(λ ₀ √ɛ_r) given in Table 1.

Table 1.

Parameter A versus the effective electrical thickness h _eff introduced in relation to equation (1)

W and L denote the width and length of the patch element and h is the geometrical thickness. For a square patch, W = L = 1.8 mm and h = 0.315 mm with the design value ɛ_r = 7.07 and λ ₀ = 10.17 mm (f = 29.5 GHz), we find A = 180 because the effective electrical thickness is about 0.0116. This means that the expected percentage bandwidth lies in the range of 2.1%. In relation to the center frequency, this corresponds to an absolute bandwidth of 620 MHz.

Attempts to reduce the effective dielectric permittivity of the substrate and thus increase the operational frequency bandwidth by creating an air cavity in the excited field volume were not pursued further due to additional technological challenges.

Further bandwidth limitations arise from the feeding topology. The individual antenna elements have to be excited with two orthogonal polarizations on two perpendicular sides. The inset feed technique avoids a bandwidth reducing line transformer, but numerical full-wave simulations result in a not feasible geometry.

The square patches were fed with λ/4-line transformers. Because of the necessity to feed the antenna ports through several layers of the 9K7 LTCC material, all potential discontinuities can be used as transformation elements, to avoid further bandwidth reductions, given that these can be suitably dimensioned through design optimization.

The multi-layer structure is depicted in Fig. 3(a) where the feed reaches from the upper plane of the patch antennas surrounded by the heat sink through the antenna ground, and then connects to a shielded stripline which is again fed to a microstrip transmission line.

Fig. 3.

Cross-section of the feedthroughs of the antenna signals of the multi-layer module mounted on a 2 mm copper heat sink, with microstrip and stripline structures on upper layers and the patch antenna array surrounded by the chamfered faces of the heat sink aperture (a) [Reference Spira, Blau, Wagner, Schulz, Gutzeit, Müller, Thomä and Hein12]. 3D view on the patch antennas with λ/4-transformers and vias for signal feed into the stripline plane (b).

It is important to note that the linear patch antenna array is actually located in a metal tub, which is formed by the antenna ground layer and the surrounding ground vias, see bottom part of Fig. 3. This design is intended to prevent any coupling of the antennas to the rest of the signal distribution network. The right-hand part of the linear array including the λ/4-microstrip line transformers and feedthroughs can be seen in detail in Fig. 4. This arrangement with the patches rotated by 45° avoids line elements between the patches and ensures that a spatially separate transmission of the orthogonal antenna signals becomes possible.

Fig. 4.

Top view of the right-hand side of the four-element dual-polarized microstrip antenna line. At the end of the λ/4-transformers, the feedthrough pads (0.25 mm in square) are clearly visible with vias in the center to the shielded stripline plane passing through the antenna ground-plane aperture (0.5 mm in square).

At the end of the λ/4-transformers, the feedthrough pads are placed together with the vias to the shielded stripline plane in their centers. This design was later integrated into the FE module, fabricated in the same LTCC technology with DuPont™ 9K7. Before integration, it was necessary to measure the return loss of each individual antenna port over the quite complex three-dimensional (3D) feed network so that there was the possibility to apply corrections as needed. In order to enable such measurements, the linear four-element array was manufactured on a separate 9K7 carrier of exactly the same layer stack as well as the antenna and feedthrough elements as the version to be integrated into the FE. The metallization around the antenna line was also identical to the design for the FE. The only difference was an additional extension of the microstrip lines to the mini-SMP plugs for characterizing the single patch antennas by vector network analysis. The fabricated LTCC carrier with the intended linear array is depicted in Fig. 5.

Fig. 5.

Left-hand panel: Top view of the separate 9K7 carrier with the four-element dual-polarized microstrip antenna line; right-hand panel: bottom view with the mini-SMP plugs with port numbers.

Antenna measurements

In a subsequent step, the reflection coefficients of the single-antenna ports were measured, in order to determine the resonant frequencies of the individual patches. The other ports were terminated with 50 Ω, and the mutual coupling was not recorded. The results are shown in Fig. 6 using the port numbers of Fig. 5. As it can be seen, the center frequencies of the four patches vary between 29.0 and 29.5 GHz, which additionally limits the usable bandwidth. The measured reflection coefficients include material and geometrical tolerances of the antenna patches, feed network, stripline-to-microstrip transition, and microstrip-to-miniSMP transition. Since the return loss was better than 20 dB at all ports, there was initially no need to modify the design of the line feeds. Alluding to Fig. 5, the geometrical differences between ports #1 and #8 and ports #4 and #5 with respect to the surrounding ground layout could be deduced approximately only.

Fig. 6.

Measured reflection coefficients of the orthogonal ports of the single patch antennas for center frequency comparison. Further details are provided in the main text.

In the last step before the integration of the antenna array into the FE, the radiation patterns of the single patches were measured in an anechoic shielded chamber. In order to emulate realistic circumstances, the antenna substrate was mounted on the intended heat sink.

The heat sink has a window slightly larger than the dimensions of the metal-free area around the four patches (31 mm × 10 mm) but with side bevels of 45° toward the antenna plane as depicted in Fig. 7. This measure should avoid the formation of a multi-mode waveguide with unfavorable effects on the radiation patterns and the polarization purity. In this context, it is to be noted that the thickness of the heat sink approaches a quarter of the free-space wavelength. From horn antennas it is well known that a large flare angle causes only a low gain increase [Reference Balanis16] but, in any case, a certain influence of this metallic environment on the antenna patterns and the polarization purity has to be expected (see below).

Fig. 7.

3D view of the chamfered faces of the 2 mm thick heat sink window. This model was used for the full-wave antenna simulation.

At this stage of development, it was only possible to measure the individual antennas and not the array pattern because this would have required an in-phase 4:1 power combiner for each polarization. The LTCC substrate with the patch antenna array was mounted on the backside of the 150 mm × 150 mm heat sink and afterward on the antenna positioner shown in Fig. 10. It is important to mention that the illumination antenna was linearly polarized at 45° with respect to the vertical direction, so that one port of the patch antennas could receive the full signal power. In this way, the maximum dynamic range of the test site can be exploited. For pattern measurements of the orthogonal polarization, the illumination antenna was rotated by 90° to the opposite direction (135°). The azimuthal patterns of the single patch antennas are depicted in Fig. 8. It must be expected from the symmetrical design and arrangement of the antenna elements that the orthogonal ports #5–#8 have the same patterns as the common ports #1–#4.

Fig. 8.

Measured azimuthal realized gain patterns of the single patches at 29.3 GHz.

A comparison of the patterns of the common ports with the patterns of the orthogonal ports (per patch element) in Fig. 8 shows indeed good agreement within the half-power beamwidth. The elevation patterns of the single elements are provided in Fig. 9. The expectation here is again that the common and orthogonal ports should reveal similar patterns to those in Fig. 8, but also an agreement between elevation and azimuth patterns is expected, at least in the range of the half-power beamwidth. This is of great importance from the point of view of polarization agility for linearly polarized received signals across the operating bandwidth; the results were presented in [Reference Spira, Thomä and Hein13].

Fig. 9.

Measured elevation patterns of the single patches at 29.3 GHz.

FE embedded antenna measurements

The analog FE with the linear antenna array embedded was eventually mounted in our antenna test range as depicted in Fig. 10. The parameters of the steerable and polarization-agile antenna array were extracted from measurements with the complete FE unit.

Fig. 10.

FE mounted on a phi positioner in the antenna test range together with a special mechanism for guiding the control, power, and LO connections [Reference Spira, Blau, Thomä and Hein14].

The FE acts as a down-converter, necessitating special provisions during acquisition of the antenna patterns. Neither a reference path nor any phase information was available from the FE. First of all, the Ludwig-III [Reference Ludwig17] co-polarization patterns were measured by synchronously rotating the linearly polarized illumination antenna and, thus, any coordinate systems transformation requiring phase values were omitted. The feed groups of the patch antenna array were oriented at ±45° relative to the x-axis in the xy-plane depicted in Fig. 7. The xz-plane denoted the horizontal plane, and the z-axis pointed toward the probe antenna. The measurements were performed with horizontal Rx polarization by combining the signals of both feed groups with equal weights. An electromagnetic full-wave simulation with AnsysEM^® software including the heat sink surrounding the antenna aperture, as depicted in Fig. 7, was performed simultaneously. In order to spread 7.1 W of thermal power of the FE, a heat sink was thermally and electrically connected to the LTCC board by a carbon layer. A miniaturization of the FE was not a major concern, therefore a thermal simulation was omitted and the heat sink size was chosen with a safe margin.

The resulting measured and simulated FE gains plotted over frequency in Fig. 11 reveal effects impacting the digital demodulation. It is clearly seen that especially the transmission line losses (patch to LNA and all other lines) in connection with the amplifier frequency response result in an inclination of the overall gain–frequency dependence. Measured and simulated horizontal patterns are shown in Fig. 12 for the selected linear phase offsets in the beamformer paths. The maximum measured offset was ±247.5°, well outside the scan range according to the usual grating-lobe criterion. This offset illustrates an array scan up to about 75°, while losing the unambiguity of the array factor but still enabling signal detection. The grating-lobe criterion [Reference Skolnik18] results in 27.52° scan angle for an array size of four isotropic radiating elements, an element spacing of 7 mm, and a frequency of 29.3 GHz. This angle is related to a phase offset of 113.76° between the radiators.

Fig. 11.

Frequency responses of the measured FE gain referred to the SMP board connector and simulated realized gain of the antenna array.

Fig. 12.

Measured (red and blue curves) and simulated (green curves) horizontal beam steering patterns for selected linear phase offsets at 29.3 GHz [Reference Spira, Blau, Thomä and Hein14].

The measured patterns agreed well with the numerical simulations, proving the expected beam steering capability. Uncalibrated phase errors of the five-bit phase shifters, as previously reported [Reference Spira, Thomä and Hein13], and amplitude imbalances give rise to deviations from the simulated responses.

Figure 13 shows scanning results for usable phase offsets, ±16° scanning was achieved with −6 dB side lobes, and about ±27° with 5 dB scan loss. A directivity of 14.9 dBi was derived from the measured half-sphere 3D gain pattern of the array steered to boresight, in excellent agreement with the numerical value of 15.1 dBi; both values were averaged to 15 dBi for the following analyses. The simulated realized gain was finally corrected to 13.2 dBi, accounting for −1.6 dB simulated antenna efficiency and ~0.2 dB of additional mismatch losses resulting from the input impedance of the LNAs. A simulation of the array with a 0.1 mm thick heat sink and otherwise unchanged parameters yielded 1.5 dB lower total realized gain and directivity compared to the 2 mm heat sink, confirming the expected impact of the heat sink aperture on the antenna properties.

Fig. 13.

Horizontal beam steering patterns of beam #1 for ±27° scan angle and 29.3 GHz, normalized to the boresight value of the 0° pattern.

Validation of the polarization purity

A special feature of the FE is the electronically adjustable polarization for investigating propagation properties and multi-user scenarios in 5G mm-wave communications. Initial measurement results concerning the polarization rotation were presented in [Reference Spira, Thomä and Hein13], revealing discrepancies compared to the intended characteristics. The simulation and measurement results were inspected and the polarization parameters of the feed groups were extracted.

Two circular polarized waves are decomposed from the complex radiating field components E_θ and E_φ [Reference Visser19]:

(2)$${\boldsymbol E}_L = \displaystyle{1 \over {\sqrt 2 }}\;( {{\boldsymbol E}_\theta -j \times {\boldsymbol E}_\varphi } ) $$

(3)$${\boldsymbol E}_R = \displaystyle{1 \over {\sqrt 2 }}\;( {{\boldsymbol E}_\theta + j \times {\boldsymbol E}_\varphi } ) \;$$

The tilt angle τ between the semi-major axis of the polarization ellipse and the E_θ unit vector, coalescing with the horizontal direction at boresight, is derived according to [Reference Visser19, Reference Hollis, Lyon and Clayton20] as:

(4)$$\tau = \displaystyle{1 \over 2}{\rm arg}\left({\;\displaystyle{{{\boldsymbol E}_L} \over {{\boldsymbol E}_R}}\;} \right)\;$$

The ratio of both semi-axes of the ellipse yields the axial ratio [Reference Visser19]:

(5)$$AR = \left\vert {\displaystyle{{\vert {{\boldsymbol E}_L} \vert + \vert {{\boldsymbol E}_R} \vert } \over {\vert {{\boldsymbol E}_L} \vert -\vert {{\boldsymbol E}_R} \vert }}} \right\vert \;$$

and in dB:

(6)$$AR_{{\rm dB}} = 20 \times {\rm lo}{\rm g}_{10}( {AR} ) \;$$

The antenna simulation delivers the E_θ and E_φ components at θ = φ = 0° for separate excitations of the two feed groups. AR and τ were then calculated with equations (5) and (4) to 2.91 and 25.5° at 29.3 GHz, respectively. Further results are listed in Table 2.

Table 2.

Simulated axial ratio and tilt angle versus frequency for 2 mm heat sink thickness

The polarization ellipses are illustrated in Fig. 14 with respect to the orientation of the antenna array. These parameters were also measured with a rotating illumination antenna in the test range at 29.3 GHz and resulted in τ = 26° and AR = 3.7, see [Reference Hollis, Lyon and Clayton20–Reference Toh, Cahill and Fusco22]. These procedures include provisions for considering an elliptical polarization of the illumination antenna.

Fig. 14.

Polarization ellipses of the feed groups at boresight with the antenna array indicated.

The polarization rotation is based on an ideal phase relationship between the two polarization components, which are weighted and summed. Figure 15 shows the measured and simulated phase differences of the FE, with large deviations from the ideal behavior caused by the 2 mm thick heat sink.

Fig. 15.

Measured and simulated phase differences between the feed groups of the FE antenna and of a dual-polarized horn antenna for indicating nearly ideal behavior. The antenna-under-test was fixed and the illumination antenna is rotated around the boresight axis.

A significant improvement can be achieved by thinning the heat sink in the vicinity of the antenna. The simulation repeated with 0.1 mm heat sink thickness resulted in AR = 17 and τ = 36.6° with a much lower residual tilt error of 8.4°. A varying phase difference subject to the polarization of an input signal cannot be compensated satisfactorily; therefore, a tuning of the heat sink geometry appears inevitable for future improved designs.

Over-the-air communication measurement setup

The applicability of the FE for 5G communications was investigated with digitally modulated broadband signals employing the setup shown in Fig. 16.

Fig. 16.

OTA digital data transmission measurement setup.

Two sets of quadrature amplitude modulated (QAM) signals, single-carrier and orthogonal frequency division multiplexed (OFDM) multi-carrier, were synthesized with the Keysight “IQtools” MATLAB^® package. The real-valued IF signal at 1.8 GHz was generated by using a Tektronix AWG7102 arbitrary waveform generator (AWG), operating at 18 GSps in interleaved output mode. A single-sideband converter ADMV1013 from Analog Devices was applied for the Tx up-conversion to a mm-wave frequency of 29.3 GHz. The dedicated evaluation board of this MMIC was operated in the IF mode in conjunction with an external 90° hybrid. The generated mm-wave signal was transmitted by a WBHDP18-40K horn antenna from Q-par Angus Ltd. with 14.5 dBi gain at 29.3 GHz. The FE as the device-under-test (DUT) was placed at an OTA distance of 1.75 m under consideration of the link budget and output back-off (BO) drive levels. The link worked in an RF laboratory, and multi-path propagation effects were involved intentionally. The DUT converts the received signals down to an IF of 1.8 GHz. A Keysight real-time oscilloscope DSOS404A with 4 GHz analog bandwidth and running 89600 VSA software was used for the signal demodulation and analysis. The FE was characterized with respect to the SMP connector on the LTCC substrate. The output signal passes an intermediate adapter board (0.7 dB loss) and an RF-cable with 4.3 dB insertion loss before entering the input of the oscilloscope. The up- and down-converters incorporate frequency quadruplers and driver amplifier circuitries for the LO frequency generation. The mixing operates with a low-side LO frequency in both cases. The Tx and Rx LO signals at a frequency of 6.875 GHz were supplied by Keysight E8257D PSG microwave signal generators without low-phase noise options. The analyzers and generators in the setup were not synchronized. In this way, the link was operated under realistic conditions.

System EVM analysis and test signals

Agile data rate adaptions are implemented in advanced digital data transmission systems. Coarse steps result from changes in the order or scheme of the complex modulation. The EVM is applied as a quality parameter, replaces the overall signal-to-noise and distortion measure, and is defined as a ratio of voltages. Prevalent definitions of the EVM differ in their normalization [Reference Vigilante, McCune and Reynaert23]. The constellation maximum is used as the reference value subsequently and the EVM is denoted by EVM_max [Reference Vigilante, McCune and Reynaert23].

Assuming mutually uncorrelated error contributions, the total EVM of a system can be approximated by the square root of the sum of all squared EVM contributions [Reference Vigilante and Reynaert24–Reference Georgiadis26]:

(7)$$EVM_{tot} = \sqrt {EVM_{AWGN}^2 + EVM_{IQ}^2 + EVM_{PN}^2 + EVM_{PA}^2 } \;$$

where EVM_AWGN comprises the thermal noise induced degradations mainly in the FE and the Tx up-converter, and also image generation issues in heterodyne concepts. EVM_IQ combines contributions by I/Q imbalances and frequency-dependent gain deviations. EVM_PN results from the integrated double sideband phase noise of the LOs. EVM_PA accounts for nonlinear distortions in the Tx output stage, and also for intermodulation effects in the receiver. Equation (7) emphasizes major but omits minor effects. The separate portions are assessed hereinafter for the OTA measurement setup, and the EVM of the FE is evaluated finally.

The signal-to-noise ratio (SNR) in the Rx and Tx chains is related to an EVM_RMS value in a simple way [Reference Vigilante and Reynaert24]. Rx IQ-impairments and the gain and phase flatness of the link were removed by the tracking and equalization of VSA software. The Tx signals were generated by means of software and an AWG. The EVM of the AWG was measured, subscripted as AWG, with impairments caused by a limited resolution and clock jitter of the involved digital-to-analog converter included. The impact of the LO phase noise on the signal quality, viz. the EVM, was investigated in the literature [Reference Kibaroglu, Sayginer, Phelps and Rebeiz10, Reference Vigilante and Reynaert24–Reference Georgiadis26]. The carrier tracking loops of the VSA fade out portions of the phase noise in addition, so that this error component is accounted for approximately. Nonlinear signal distortions, incorporated as EVM_PA in (7), were prevented by operating all parts of the link with an adequate BO from the compressed output power. The BO encompasses a safety margin and the peak-to-average power ratio (PAPR) of the transmitted signal.

Test signals

The 16-, 64-, and 256-QAM signals with a bandwidth of 400 MHz were generated as single- and multi-carrier waveforms, reflecting general requirements of the wireless IEEE 802.11 and mm-wave 5G NR applications. SC-QAM signals with a symbol rate of 400 Mbaud were generated. A pulse shaping filter with root-raised cosine characteristic and a roll-off factor of 0.13 was utilized. The PAPR for identifying the required BO was calculated from the oversampled waveform, up-converted to an IF of 1.8 GHz, to 7 ± 0.4 dB for all three QAM signals. The EVM values of the AWG measured at the input of the 90°-hybrid, see Fig. 16, are indicated in Table 4. The power level for 64-QAM was −20.1 dBm.

OFDM signals with 16-/64-/256-QAM data carrier modulation were synthesized according to the IEEE 802.11a standard and modified to account for 5G NR enhancements. The baseband specifications and data rates calculated according to [Reference Goldsmith27] are given in Table 3. Four preamble, one signal and 25 data symbols constitute one OFDM burst. The preamble and pilot subcarriers were randomized for equalizing the burst power envelope and the channel spectrum.

Table 3.

OFDM signal baseband specifications and calculated data rates at 400 MHz channel bandwidth

The EVM values of the AWG, as measured as before, are listed in Table 4 as EVM_max values [Reference Vigilante, McCune and Reynaert23] for a comparison to SC-QAM results. The data EVM dominated in our case and all data carriers were identically modulated. The EVM_rms values, more common for OFDM, are replaced by EVM_max in the table. The power level was −18.9 dBm for 64-QAM, since the AWG IF signal attenuation was 4 dB less. The calculated PAPRs of the OFDM signals were 10.1 ± 0.1 dB.

Table 4.

Measured AWG and 1.75 m link EVM values and calculated data rates at 400 MHz modulation bandwidth

1.75 m link measurements

The main contributions to the total EVM, as named before, were analyzed for the SC-64-QAM test in this section for a link budget and the EVM of the FE was extracted. The dynamic range of the link was derived from these results finally.

Total EVM

The total link EVM was measured to 1.63% and consists of 37.2 dB SNR of the FE (0.9% EVM), 0.53% AWG EVM (see Table 4), 44 dB up-converter SNR (0.41% EVM), and two times 0.835% EVM_PN for the phase noise of the two LO generators, viz. 1.63 = √ (0.9² + 0.53² + 0.41² + 0.835² + 0.835²).

The presented FE is characterized by its SNR, related to the NF and bandwidth, and the LO phase noise induced degradation and shows in this test a state-of-the-art performance of 1.23% EVM, de-embedded from 1.63% total EVM. The results of the link measurements of all generated signals are summarized in Table 4 together with the corresponding data rates.

The 64-QAM OFDM test was repeated with an 800 MHz system frequency and resulted in 2.6% total EVM at a data rate of 3.4 Gb/s. A screenshot of the VSA displaying the measured constellation diagram for SC 64-QAM is shown in Fig. 17.

Fig. 17.

VSA screenshot of an SC 64-QAM measurement with 300 symbols. The constellation diagram is given in the upper left subplot.