Introduction
To meet the stringent demands of emerging multi-standard wireless-communication systems, a vast research effort in the key components of their RF front-ends – e.g., multi-band bandpass filters (BPFs) with high-selectivity responses through multi-transmission-zero (TZ) creation – is being carried out [Reference Crnojevic-Bengin1]. In order to produce a plurality of stopband TZs so as to attain very-abrupt filtering cut-off slopes, planar/multilayer dual-band BPFs employing transversal signal-interference topologies and multiple signal-transmission paths have been reported [Reference Gómez-García, Sánchez-Renedo, Jarry, Lintignat and Barelaud2–Reference Yang, Malki and Gómez-García16]. Planar multi-TZ BPFs with spectrally symmetrical and asymmetrical dual passbands using transversal signal-interference filtering sections were firstly presented in [Reference Gómez-García, Sánchez-Renedo, Jarry, Lintignat and Barelaud2] and [Reference Gómez-García, Muñoz-Ferreras and Sánchez-Renedo3], respectively. Similarly, an asymmetrical dual-band BPF with self-coupled microstrip sections and a low-pass-type filtering function was studied in [Reference Choudhary and Chaudhary4]. In addition, through a dual-mode slotted circular microstrip patch resonator, a second-order dual-band BPF with five TZs was discussed in [Reference Zhang, Zhu and Luo5]. To further reduce circuit size, sharp-rejection dual-band BPFs based on substrate-integrated-waveguide (SIW) multi-mode resonators in [Reference Yang, Zhu and Wang6] and [Reference Zhu and Dong7] and complementary-split-ring-resonator-loaded SIW eighth-mode resonators in [Reference Liu, Li, Qin, Yin and Qiu8] were engineered. Nevertheless, all these developed prior-art dual-band BPFs have limited lower/upper stopband ranges due to their intrinsic DC transmission and their frequency-periodic behavior in some cases. Thus, in order to attain extended stopband ranges and high-selectivity responses with multiple out-of-band TZs simultaneously, multi-signal-transmission-path-based BPFs with frequency-symmetrical and asymmetrical dual passbands that, respectively, exploit coupled-line-fed single-/multi-mode resonators in [Reference Zhang and Zhu9–Reference Yang, Malki, Muñoz-Ferreras and Gómez-García13] and resonator-based structures in [Reference Xu, Ma and Du14, Reference Fu, Li, Liu, Cheng and Qiu15] were explored. Moreover, by introducing several embedded notched bands and TZs, a family of dual-/multi-band BPFs were synthetically designed with in-series-cascaded and coupled resonators in [Reference Gómez-García, Yang and Psychogiou17–Reference Wu, Ma and Wang19], respectively. However, none of the above dual-band BPFs exhibits the three following desired features at the same time: (i) high-order passbands, (ii) large number of inter-band TZs with extended lower/upper stopbands, and (iii) increased out-of-band power-attenuation levels.
In this paper, a type of two-layer signal-interference (i.e., multi-RF-signal-propagation-path) fourth-order high-selectivity dual-band BPF with multiple TZs is reported. Through two vertically coupled microstrip-to-slotline vertical transitions and in-parallel asymmetrical microstrip lines arranged in a closed loop, a class of second-order BPFs with spectrally symmetrical/asymmetrical dual passbands and three inter-band TZs is firstly discussed. By adding two shunted stepped-impedance microstrip stubs at the input and output accesses of the previous frequency-symmetrical dual-band BPF, two more transmission poles for each passband along with two additional TZs at the lower and upper stopband regions can be obtained. It yields a fourth-order dual-band BPF with sharper cut-off slopes and higher out-of-band power-rejection levels when compared to the preliminary design presented in [Reference Yang, Malki and Gómez-García16]. In the “Multilayer signal-interference dual-band BPFs” section, the operational principles of these devised BPFs are detailed. Finally, in the “Experimental results and discussions” section, a fourth-order dual-band BPF prototype with multiple stopband TZs is developed and tested.
Multilayer signal-interference dual-band BPFs
The layout of the conceived fourth-order dual-band BPF in a two-layer substrate is shown in Fig. 1. As observed, it consists of two parts: (i) two microstrip-to-slotline vertical transitions loaded with two shunted Y-shaped stepped-impedance microstrip stubs at the input/output ports on the top layer and (ii) a signal-interference in-parallel asymmetrical microstrip-line-based closed loop printed on the bottom layer. Through the couplings with the microstrip closed loop, these two microstrip-to-slotline transitions are back-to-back cascaded. To mitigate unwanted radiation effects, two identical shunted high-impedance microstrip lines instead of low-impedance ones for the Y-shaped stubs are used. Meanwhile, compared with the folded slotline resonators reported in [Reference Yang, Malki and Gómez-García16], two meandered slotline resonators with folded slotline segments at their short-circuited ends are herein utilized to avoid the unexpected coupling effect between the slotline segments and the shortest path of the microstrip closed loop along with the high-impedance microstrip lines of the Y-shaped stubs, so as to attain a satisfactory dual-band BPF response. In this context, the RF properties of the proposed filter can be determined with its associated transmission-line (TL) equivalent circuit [Reference Yang, Zhu, Zhang, Wang, Choi, Tam and Gómez-García20]. As illustrated in Fig. 2, the open-circuit-ended microstrip resonators, the short-circuit-ended meandered slotline resonators, the shunt open-circuit-ended Y-shaped stepped-impedance microstrip stubs, and the in-parallel asymmetrical microstrip-line-based microstrip closed loop of the dual-band BPF in Fig. 1 are correspondingly replaced by the open-circuit-ended stubs, the short-circuit-ended stubs, two shunt open-circuit-ended two-section stubs, and the in-parallel asymmetrical cascaded TL segments, respectively. Their design impedances are set as Z m, Z s, Z m1, Z m2, Z m3, and Z m4, while their associated electrical lengths are θ, θ 1, and θ 2. Meanwhile, Z 0 is the 50-Ω feeding line impedance, and five pairs of transformers with various turns ratios of N m, N s, N m1, and N m2 are used to quantitatively model the impedance variations of the coupled microstrip and slotline segments [Reference Yang and Gómez-García21].
3D layout of the proposed fourth-order high-selectivity dual-band BPF with multiple TZs composed of (i) two microstrip-to-slotline vertical transitions with two shunted Y-shaped stepped-impedance microstrip lines loaded at the input and output accesses on the top layer and (ii) a transversal signal-interference microstrip closed loop on the bottom layer.
TL-based equivalent circuit of the proposed high-selectivity fourth-order dual-band BPF with multiple TZs corresponding to the layout shown in Fig. 1.
Second-order frequency-symmetrical dual-band BPF
The TL equivalent circuit is set as lossless with N m = N s = N m1 = N m2 = 1, and the electrical lengths are selected as θ 1 = θ = 90° and θ 2 = 2θ = 180° at the design frequency f D. By ignoring the shunt open-circuit-ended two-section stubs loaded at the input and output ports of the engineered fourth-order dual-band BPF, a second-order dual-band BPF with two frequency-symmetrical passbands is firstly discussed. Based on the ABCD matrix of its associated equivalent TL circuit network, a desired second-order BPF response is derived. Due to the employed in-parallel-connected asymmetrical TLs, three inter-band TZs are produced at the following spectral locations:
\begin{equation}{f_{{\text{TZ}}1}} = \frac{{2{f_{\text{D}}}}}{\pi }{\tan ^{ - 1}}\sqrt {\frac{{2{z_{{\text{m3}}}} + {z_{{\text{m4}}}}}}{{2{z_{{\text{m3}}}} - {z_{{\text{m4}}}}}}} \end{equation}
\begin{equation}{f_{{\text{TZ}}2}} = {f_{\text{D}}}\end{equation}
\begin{equation}{f_{{\text{TZ}}3}} = 2{f_{\text{D}}} - \frac{{2{f_{\text{D}}}}}{\pi }{\tan ^{ - 1}}\sqrt {\frac{{2{z_{{\text{m}}3}} + {z_{{\text{m}}4}}}}{{2{z_{{\text{m}}3}} - {z_{{\text{m}}4}}}}} \end{equation}where z m3 and z m4 are normalized-impedance parameters. By assuming z m3 = z m4 and θ 1 = θ 2 = θ = 90°, a fifth-order wideband BPF with specified in-band equi-ripple level of 0.043 dB and fractional bandwidth (FBW) of 105% is initially determined [Reference Yang, Zhu, Zhang, Wang, Choi, Tam and Gómez-García20]. Then, in order to make the 3-dB cut-off frequencies of the first passband at the lower region and the second passband at the upper region equal to the ones of the fifth-order wideband BPF, the design impedances of the proposed second-order BPF with inter-band-TZ FBW of 51.5% (defined as [f TZ3 − f TZ1]/f D × 100%), minimum inter-band power-rejection level of 20 dB, and impedance ratio R = z m4/z m3 = 1.3777 are determined. Figure 3 plots the associated frequency responses. As observed, its two passbands are centered at 0.5725f D and 1.4275f D with minimum in-band power-matching levels of 22.585 dB, 3-dB absolute bandwidths of 0.195f D, 20-dB minimum inter-band power-suppression level, and being symmetrically placed with regard to the TZ at f D. In addition, various examples of frequency responses of this spectrally symmetrical dual-band BPF for different inter-band-TZ FBWs and R values are represented in Fig. 4. In all cases, the in-band power-matching levels of the two passbands are kept higher than 22.585 dB and the minimum inter-band power-suppression level is below 20 dB. As shown in Fig. 4, the absolute bandwidths of the two passbands become larger, while the inter-band-TZ FBW is reduced as the R value is increased.
Theoretical power transmission (|S 21|) and reflection (|S 11|) responses of the proposed second-order spectrally symmetrical dual-band BPF with inter-dual-passband-TZ FBW of 51.5% and minimum inter-band power-rejection level of 20 dB (normalized design impedances: z m = 0.5536, z s = 1.4324, z m3 = 0.9176, and z m4 = 1.2642) and a comparative fifth-order wideband BPF specified with in-band equi-ripple level of 0.043 dB (i.e., minimum in-band power-matching level of 20.06 dB) and FBW of 105% (normalized design impedances: z m = 1.0922, z s = 1.5884, and z m3 = z m4 = 0.7446).
Theoretical power transmission (|S 21|) and reflection (|S 11|) responses of the proposed second-order frequency-symmetrical dual-band BPF with different inter-dual-passband TZ FBWs of 54.8%, 51.5%, and 47.2% along with their associated r values of 1.3056, 1.3777, and 1.4731, but with the same minimum inter-band power-rejection level of 20 dB and minimum in-band power-matching levels of 22.585 dB (case I: z m = 0.3696, z s = 1.4324, z m3 = 1.3182, and z m4 = 1.721; case II: z m = 0.5536, z s = 1.4324, z m3 = 0.9176, and z m4 = 1.2642; and case III: z m = 0.3716, z s = 1.933, z m3 = 0.5132, and z m4 = 0.756).
Second-order frequency-asymmetrical dual-band BPF
In order to demonstrate the design flexibility of the devised second-order dual-band BPF, its counterpart allowing to obtain two frequency-asymmetrical passbands and three TZs at the inter-band region is proposed. As illustrated in Fig. 5(a), the longest path of the in-parallel-connected TL closed loop shown in Fig. 2 is reshaped by two in-series-cascaded replicas of stepped-impedance TL segments. The design impedances and electrical lengths of the employed stepped-impedance TL segments are set as Z m41 and Z m42 and θ 21 and θ 22, respectively. Following the operational principle of the signal-interference stepped-impedance-line dual-band BPF reported in [Reference Gómez-García, Muñoz-Ferreras and Sánchez-Renedo3], the conceived second-order spectrally asymmetrical dual-band BPF is herein designed with normalized impedances: z m = 0.7002, z s = 1.7824, z m3 = 0.652, z m41 = 1.086, and z m42 = 1.734, and electrical lengths: θ 1 = θ = 90°, θ 21 = 65°, and θ 22 = 30° at f D. Figure 5(b) plots its second-order frequency-asymmetrical dual-band BPF response. Specifically, two passbands are centered at 0.59f D and 1.5775f D and exhibit 3-dB absolute bandwidths of 0.35f D and 0.125f D (i.e. 2.8:1 ratio), respectively, and three inter-band TZs are produced at 0.865f D, 1.11f D, and 1.395f D.
Proposed second-order frequency-asymmetrical dual-band BPF. (a) Conceived stepped-impedance TL-based circuit acting as the longest path of the transversal signal-interference in-parallel-connected TL closed loop in Fig. 2. (b) Theoretical power transmission (|S 21|) and reflection (|S 11|) responses of the designed second-order dual-band BPF with two frequency-asymmetrical passbands and three inter-band TZs (normalized design impedances: z m = 0.7002, z s = 1.7824, z m3 = 0.652, z m41 = 1.086, and z m42 = 1.734).
Fourth-order frequency-symmetrical dual-band BPF
Although second-order multi-TZ dual-band BPFs with two frequency-symmetrical and asymmetrical passbands have been designed, the out-of-band selectivity of their two passbands is still poor as no close-to-passbands TZs are produced at the lower and upper stopband regions. Besides, their lower-order dual-band BPF responses result in relatively low inter-band power-attenuation levels. In order to address these drawbacks, a high-selectivity fourth-order dual-band BPF that builds upon the previous second-order frequency-symmetrical dual-band BPF block with two shunt open-circuit-ended two-section stubs loaded at the input and output ports is proposed, as depicted in Fig. 2. Through this pair of shunted two-section stubs, two more transmission poles for each passband and a pair of additional close-to-passband TZs can be attained [Reference Gómez-García, Yang, Muñoz-Ferreras and Psychogiou22]. Herein, the frequency locations of the two TZs associated with these two-section stubs can be formulated as follows:
\begin{equation}{f_{{\text{TZ}}4}} = \frac{{2{f_{\text{D}}}}}{\pi }{\tan ^{ - 1}}\sqrt {\frac{{{z_{{\text{m}}2}}}}{{{z_{m1}}}}} \end{equation}
\begin{equation}{f_{{\text{TZ}}5}} = 2{f_{\text{D}}} - \frac{{2{f_{\text{D}}}}}{\pi }{\tan ^{ - 1}}\sqrt {\frac{{{z_{{\text{m}}2}}}}{{{z_{{\text{m}}1}}}}} \end{equation}where z m1 and z m2 are normalized-impedance parameters. Thus, the overall filter combines the three inter-band TZs created by the in-parallel-connected closed loop as given by Eqs (1)–(3) and the two TZs associated with the shunted two-section stubs as detailed in Eqs (4) and (5), which can be independently controlled. In this manner, in order to attain the intended sharp-rejection dual-band BPF with multiple close-to-passband TZs, the derived two additional close-to-passband TZs (i.e., TZ4 and TZ5) are expected to be located within the spectral ranges (0, 0.5f D) and (1.5f D, 2f D), respectively, which yields z m2/z m1 < 1. Following an analogous design procedure as for the discussed second-order frequency-symmetrical dual-band BPF and based on Eqs (1)–(5), the design impedances of the fourth-order dual-band BPF for an inter-band-TZ FBW of 34% with R = 1.7228 and an out-of-band-TZ FBW of 140% (defined as [f TZ5− f TZ4]/f D× 100%) with R 1 = z m2/z m1 = 0.2589 are determined. Figure 6 depicts the frequency responses of the derived fourth-order dual-band BPF. As can be seen, when compared to the second-order dual-band BPF block, the fourth-order dual-band BPF in Fig. 2 has deeper inter-band power-rejection levels and sharper filtering cut-off slopes. Specifically, its two passbands are centered at 0.5814f D and 1.4186f D and both exhibit a 0.255f D absolute bandwidth.
Theoretical power transmission (|S 21|) and reflection (|S 11|) responses of the proposed high-selectivity fourth-order frequency-symmetrical dual-band BPF in Fig. 2 (normalized design impedances: z m = 1.356, z s = 1.322, z m1 = 1.638, z m2 = 0.424, z m3 = 0.938, and z m4 = 1.616).
Fourth-order frequency-asymmetrical dual-band BPF
As in the lower-order dual-band BPF counterpart, high-order dual-band BPFs with frequency-asymmetrical responses can be realized. For illustration purposes, Fig. 7(a) shows the TL equivalent circuit of the conceived high-order frequency-asymmetrical dual-band BPF. Herein, compared with the discussed second-order frequency-asymmetrical dual-band filter in Fig. 5, two identical open-ended multi-segment stepped-impedance stubs are introduced at the input and output accesses, and the longest path of the transversal signal-interference in-parallel-connected TL closed loop is also modified by using two back-to-back cascaded three-segment-based TLs with different impedances and electrical lengths. Through the carefully selected design impedance values and the associated electrical lengths, the devised frequency-asymmetrical dual-band BPF can attain a high-order dual-passband response with remarkable in-band performance and multiple stopband TZs as exemplified in Fig. 7(b).
Proposed fourth-order frequency-asymmetrical dual-band BPF. (a) Conceived transmission-line equivalent circuit with z m = 58.5 Ω, z m1 = 102.2 Ω, z m2 = 85.2 Ω, z m3 = 32.7 Ω, z m41 = 54.6 Ω, z m42 = 90.5 Ω, z m43 = 53.7 Ω, z s = 87 Ω, θ = θ 1 = 90°, θ 21 = 61.7°, θ 22 = 33.2°, θ 23 = 59.7°, θ 31 = 30.5°, and θ 32 = 71.2°. (b) Theoretical power transmission (|S 21|) and reflection (|S 11|) responses.
Experimental results and discussions
To verify the experimental feasibility of the above-discussed high-selectivity fourth-order frequency-symmetrical dual-band BPF, a proof-of-concept microstrip prototype based on the conceived two-layer layout in Fig. 1 is designed, simulated, and tested (f D = 2 GHz). It is manufactured using Rogers RO4003C substrate with relative dielectric constant ε r = 3.55, dielectric thickness h = 0.813 mm, metal thickness t = 35 µm, and dielectric loss tangent tan(δ D) = 0.0027. Figure 8 depicts the photographs of the constructed two-layer dual-band BPF prototype through the views of different layers (i.e., top, bottom, and middle ones), and the total circuit size of the manufactured BPF prototype is 135 × 90 mm2. The theoretical, electromagnetically (EM) simulated, and measured results of this developed fourth-order dual-band BPF are compared in Fig. 9(a). As shown, a fairly close agreement among these three sets of frequency responses is obtained. The theoretical responses of the fourth-order dual-band BPF in Fig. 6 were initially calculated with Z m = 67.8 Ω, Z s = 66.1 Ω, Z m3 = 46.9 Ω, Z m4 = 80.8 Ω, Z m1 = 81.9 Ω, and Z m2 = 21.2 Ω, respectively. However, in order to make the EM-simulated responses fairly match the theoretical ones, the implemented design impedances of the proposed dual-band BPF during the EM simulation were adjusted as 
${Z{^{\prime}}_{\text{m}}}$ = 66.815 Ω, 
${Z{^{\prime}}_{{\text{s1}}}}$ = 92.294 Ω, 
${Z{^{\prime}}_{{\text{m3}}}}$ = 50.144 Ω, 
${Z{^{\prime}}_{{\text{m4}}}}$ = 77.602 Ω, and Z m1 = 81.965 Ω, and the mathematically equivalent Z m2 = 17.47 Ω was realized with two shunted 34.939-Ω quarter-wavelength microstrip lines. Through these two sets of impedance values, the turns ratios of the utilized transformers can be extracted as N 1 = 1.007, N 2 = 0.846, N 3 = 0.967, and N 4 = 1.02, respectively.
Photographs of the manufactured two-layer microstrip prototype of high-selectivity fourth-order frequency-symmetrical dual-band BPF corresponding to the layout in Fig. 1 (dimensions: l in = l out = 45, L 1 = 21.68, L 2 = 22.27, L 3 = 22.9, L 4 = 9.07, L 5 = 3.74, L 6 = 10.65, L 7 = 29.98, L 8 = 33.79, L 9 = 6.65, L 10 = 32.8, w in = w out = 1.82, W 1 = 1.062, W 2 = 0.69, W 3 = 3.06, W 4 = 0.43, W 5 = 0.78, W 6 = 1.77 [unit: mm], α 1 = 45°, and α 2 = 90°]. (a) Top-layer view. (b) Bottom-layer view. (c) Middle-layer view.
Frequency responses of the manufactured two-layer microstrip prototype of high-selectivity fourth-order frequency-symmetrical dual-band BPF in Fig. 8. (a) Theoretical, em-simulated, and measured power transmission (|S 21|) and reflection (|S 11) responses. (b) Em-simulated and measured in-band group-delay (τ g) responses.
The main measured performance metrics of the two passbands of the built fourth-order dual-band BPF microstrip prototype are listed as follows: center frequencies of 1.154 GHz and 2.818 GHz, minimum in-band power-insertion-loss levels of 0.43 dB and 0.92 dB, in-band power-matching levels of above 22.6 dB and 18.47 dB, and 3-dB absolute bandwidths of 451.44 MHz and 563.04 MHz (i.e., 39.12% and 19.98% in fractional terms). The inter-band power-rejection levels are above 28.68 dB and the stopband power-attenuation levels are higher than 40.92 dB from DC to 4.64 GHz (i.e., without considering the passband and inter-band regions). With regard to the minor discrepancies among the bandwidths of the EM-simulated and measured passbands, they are mainly attributed to the radiation and impedance-dispersion effects of the meandered slotline resonators utilized in the practical implementation. In addition, the EM-simulated and measured in-band group-delay responses of these two passbands are depicted in Fig. 9(b). The measured maximum in-band group-delay variations are 1.59 ns and 2.04 ns for the lower and upper passbands, respectively. Moreover, in order to explain the discrepancies between the EM simulated and measured results in Fig. 9(a), a tolerance analysis of the simulated frequency responses of the proposed dual-band BPF against the machining tolerance of ±0.03 mm is provided, as depicted in Fig. 10. As can be seen, the dual-passband responses of the proposed BPF are almost unchanged, whereas the in-band power-matching levels associated with the machining tolerance of +0.03 and −0.03 mm, respectively, are slightly degraded, especially for the second passband, but are still better than 16.3 dB. Furthermore, a comparison between the proposed fourth-order dual-band BPF and other prior-art dual-band BPFs is detailed in Table 1. As can be seen, although the manufactured high-selectivity fourth-order dual-band BPF prototype of this work occupies a larger circuit size, it features superior performance in terms of higher number of out-of-band TZs, relatively wider FBWs, lower minimum in-band power-insertion-loss levels, and higher stopband power-attenuation levels. The development of further-miniaturized multi-TZ high-order BPFs using a multilayer substrate is left as future research work to be explored.
Comparison of the simulated power transmission (|S 21|) and reflection (|S 11|) responses (i.e., ideal case) in Fig. 9(a) and two cases attributed to all the dimensions with a machining tolerance of ±0.03 mm for the high-selectivity fourth-order frequency-symmetrical dual-band BPF in Fig. 8.
Comparison with other prior-art high-selectivity dual-band BPFs
* CFs: center frequencies; FO: filter order; IPLs: inter-band power-rejection levels; SPLs: stopband power-attenuation levels; *: estimated from graphs, λ g: guided wavelength at the center frequency of the first passband.
Conclusion
The development of a class of two-layer dual-band BPFs with multiple inter-band and out-of-band TZs has been reported. Second-order dual-band BPFs featured with frequency-symmetrical and asymmetrical passbands along with three TZs at the inter-band region have been firstly discussed. To further increase the filter order of each passband and the stopband power-attenuation levels with more close-to-passband TZs, a fourth-order high-selectivity dual-band BPF has been then developed. Their operational principles using the associated TL equivalent circuits have been described in detail. Finally, a two-layer fourth-order dual-band BPF microstrip prototype has been manufactured and tested for practical-validation purposes.
Funding statement
This work was supported in part by the Spanish Ministry of Economy, Industry, and Competitiveness (State Research Agency) under Project PID2023-149096OB-I00 and in part by SYDNICON PTY LTD under project PC-CONT-2023-137.
Competing interests
The authors report no conflict of interest.
Li Yang received the M.Sc. degree in electrical and electronics engineering and Ph.D. degree in electrical and computer engineering from the University of Macau (UM), Macau SAR, China, in 2013 and 2018, respectively. In 2018, he was a Postdoctoral Fellow with the Department of Electrical and Computer Engineering, UM. Since 2018, he joined as a Postdoctoral Researcher with the Department of Signal Theory and Communications, University of Alcalá, Alcalá de Henares, Spain, where he worked as a GOT ENERGY TALENT (GET)-COFUND MSCA Fellow during 2020–2022, and now he is a Senior Research Scientist. His current research interests include the synthesis and design of RF/microwave planar and multilayered passive filters, reconfigurable filters, reflectionless filters, filtering antennas, and multifunctional circuits and systems. Until now, he has authored/coauthored 62 papers in international journals and 45 papers in international conferences.
Dr. Yang serves as a Technical Reviewer for several prestigious IEEE journals (e.g., TMTT, MWCL, TCAS-I, and TCAS-II) and a Technical Program Committee Member for several IEEE and EuMA conferences. He is also a Young-Professional Affiliate Member of the IEEE MTT-S Microwave Passive Components and Transmission Line Structures (MTT-4) and the IEEE MTT-S Filters (MTT-5) Technical Committees. He is currently an Associate Editor of IEEE Transactions on Circuits and Systems II: Express Briefs and IEEE Access. He was a recipient of the Best Student Paper Award of the International Symposium on Antennas and Propagation 2010 in Macau and the Second Prize of the Macau Natural Science Award in 2020.
Mohamed Malki received the B.Sc. degree in telecommunication engineering, the M.Sc. degree in telecommunication engineering in the specialized area of Space and Defense Technologies, and a Ph.D. degree in electrical engineering (with honors) from the University of Alcalá, Madrid, in 2020, 2022, and 2024, respectively. In 2022, he joined the Department of Signal Theory and Communications at the University of Alcalá, Madrid, Spain, as a Research Assistant. He was a Visiting Researcher with the Emerging Device Technology (EDT) Research Lab at the University of Birmingham, UK, in 2023. His main research interests include the analysis, design, and construction of microwave filters and multiplexers.
Dr. Malki was the recipient of the 2023 HISPASAT award for the Best M.Sc. Thesis in New Technologies for Communication Satellites, awarded by the Spanish Official College of Telecommunication Engineers (COIT/AEIT), the 2023 European Microwave Association (EuMA) Internship Award, and the 1st Best Paper Award at the 2024 IEEE International Microwave and Antennas Symposium (IMAS).
Xi (Forest) Zhu received the B.Eng. and Ph.D. degrees in electronic engineering from the University of Hertfordshire, Hatfield, UK, in 2005 and 2008, respectively. Since 2016, he has been with the School of Electrical and Data Engineering, University of Technology Sydney (UTS), NSW, Australia. He has authored or coauthored more than 150 papers in international journals and conferences. His current research interests primarily focus on the design of analog/mixed-signal integrated circuits and radio frequency integrated circuits (RFICs) for wireless communication and radar sensing applications. In 2023, he received the prestigious ARC Future Fellowship. Furthermore, in both 2022 and 2023, Dr. Zhu was named Australia’s top researcher in the field of Microelectronics and Electronic Packaging by The Australian Research Magazine.
Dr. Zhu is a Senior Associate Editor for the IEEE Transactions on Circuits and Systems I: Regular Papers and he was an Associate Editor for the IEEE Transactions on Circuits and Systems II: Express Briefs between 2022 and 2024. Also, he serves as a member of the Technical Review Board for several IEEE journals and conferences, including ISCAS which is the flagship conference organized by the IEEE CAS-S. He is also a member of the IEEE CAS-S Analog Signal Processing Technical Committees (ASPTC).
Roberto Gómez-García received the Telecommunication Engineer and Ph. D. degrees from the Polytechnic University of Madrid, Madrid, Spain, in 2001 and 2006, respectively. He is currently a Full Professor at the Department of Signal Theory and Communications, University of Alcalá, Alcalá de Henares, Spain. His current research interests include the design of fixed/tunable high-frequency filters and multiplexers in planar, hybrid, and monolithic microwave-integrated circuit technologies; multifunction circuits and systems; software-defined radio and radar architectures for telecommunications, remote sensing, and biomedical applications; and displacement RF sensors. In these topics, he has authored/coauthored about 150 papers in international journals and 180 papers in international conferences. Dr. Gómez-García serves as a member of the Technical Review Board for several IEEE and EuMA conferences. He is also a member of the IEEE MTT-S Filters (MTT-5), the IEEE MTT-S RF MEMS and Microwave Acoustics (MTT-6), the IEEE MTT-S Wireless Communications (MTT-23), the IEEE MTT-S Biological Effects and Medical Applications of RF and Microwave (MTT-28), and the IEEE CAS-S Analog Signal Processing Technical Committees. He was a recipient of the 2016 IEEE Microwave Theory and Techniques Society (MTT-S) Outstanding Young Engineer Award. He was an IEEE CAS-S Distinguished Lecturer (2020–2022). He was an Associate Editor of the IEEE Transactions on Microwave Theory and Techniques from 2012 to 2016, the IEEE Transactions on Circuits and Systems I: Regular Papers from 2012 to 2015, the IEEE Microwave and Wireless Components Letters from 2018 to 2020, and other journals such as IEEE Access, IET Microwaves, Antennas, and Propagation, the International Journal of Microwave and Wireless Technologies, and the IEEE Journal of Electromagnetics, RF and Microwaves in Medicine and Biology. He was a Senior Editor of the IEEE Journal on Emerging and Selected Topics in Circuits and Systems from 2016 to 2017 and the MTT-S Newsletter Working Group Chair from 2019 to 2021. He was a Guest Editor for several Special/Focus Issues and Sections in IEEE and IET journals. He was the Editor-in-Chief of the IEEE Microwave and Wireless Technology Letters. Currently, he is a Senior Associate Editor of the IEEE Transactions on Circuits and Systems I: Regular Papers, TC-5 Topic Editor of the IEEE Journal of Microwaves, and a Track Editor member of Electromagnetic Science. He is also an elected member of the MTT-S Administrative Committee.
Open access
